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Measuring nanoamperes

MEASURING LOW CURRENTS CAN BE TRICKY. CLEVER ANALOG- DESIGN TECHNIQUES AND THE RIGHT PARTS AND EQUIPMENT CAN HELP.

Thousands of applications require a circuit to measure a small current. One of the most common tasks is the measurement of photodiode current to infer the light impinging on the diode. Scientific applications, such as CT (computer-tomography) scanners, gas chromatographs, and photo-multiplier and particle and beam monitoring, all require low-level current measurements. In addition to these direct applications, the manufacturers of semiconductors, sensors, and even wires must measure extraordinarily low currents to characterize their devices. Leakage current, insulation-resistance measurements, and other parameters require consistent, accurate measurementsto establish data-sheet specifications.

Few engineers realize, however, that the data sheet of a part is a contractual document. It specifies the behavior of the device, and any disputes over the operation of the part always come down to the specs on the data sheet. Recently, a customer of a large analog-IC company threatened legal action against the manufacturer, claiming that the parts he had purchased exhibited far higher operating currents than the submicroampere levels that the company specified. It turned out that the PCB (printed-circuit-board)- assembly house was properly washing the board but that assemblers were picking up the PCB and leaving fingerprints on a critical node. Because it could measure these tiny currents, the semiconductorcompany proved that its parts were working correctly; the leakage currentwas due to dirty PCBs.

The difficulty with measuring small currents is that all kinds of other effects interfere with the measurement (see sidebar “History of current measurements” at www.edn.com/070426cs). This article looks at two breadboard circuits that must handle surface leakage, amplifier- bias-current-induced errors, and even cosmic rays. As in almost all circuits, EMI (electromagnetic interference) or RFI (radio-frequency interference) can induce errors, but, at these low levels, even electrostatic coupling can cause a problem. As the currents you measure drop into the femtoampere range, the circuits are subject toeven more interfering effects. Humidity changes the value of capacitors andcauses higher surface leakage. Vibrationsinduce piezoelectric effects in thecircuit. Minor temperature variations,even from a room fan, cause temperaturegradients in the PCB that give falsereadings. Even room light can degradethe accuracy of measurements; lightfrom fluorescent fixtures can enter theglass ends of a detector diode and causeinterference (Reference 1).

Small currents require accurate measurement if you want to characterize the performance of quartz-crystal oscillators. Jim Williams, a staff scientist at Linear Technology and longtime EDN contributor, shares a circuit he designed for a customer who needed to measure the rms current in a 32-kHz watch crystal (Figure 1). One difficulty with this measurement is that even an FET probe’s 1-pF loading can affect the crystal oscillation. Indeed, one of the goals of current measurement is to establish the sizing of the low-value capacitor you use with every crystal oscillator. A further difficulty of this measurement is that it must measure accurately and in real time at 32 kHz, which rules out the use of an integrating capacitor. The signal is a complex ac signal that the system designer mustconvert to an rms value for evaluation.

AT A GLANCE

Physics and noise limit the measurement of small currents. Early mechanical meters could resolve femtoamperes. JFET and CMOS amplifiers are suitable for measurements. To measure femtoampere-level currents, integrate the current into a capacitor. Integrated parts can measure femtoamperes and provide 20-bit outputs.

“Quartz-crystal rms operating current is critical to long-term stability, temperature coefficient, and reliability,” says Williams. The necessity of minimizing introduced parasitics, especially capacitance, complicates accurate determination of rms-crystal current, especially in micropower-crystal types, he says. Figure 2’s high-gain low-noise amplifier, he explains, combines with a commercially available closed-core current probe to permit the measurement, and an rmsto- dc converter supplies the rms value. The dashed lines indicate a quartz-crystal test circuit that exemplifies a typical measurement situation. Williams uses a Tektronix CT-1 current probe to monitor crystal current and introduce minimal parasitic loading. A coaxial cable feeds the probe’s 50
output to A1; A1 and A2 take a closed-loop gain of 1120, and the excess gain over a nominal gain of 1000 corrects for the CT-1’s 12% lowfrequencygain error at 32.768 kHz.

Williams investigates the validity ofthis gain-error correction at one sinusoidal frequency—32.768 kHz—with aseven-sample group of Tektronix CT-1s.He reports that device outputs are collectivelywithin 0.5% of 12% down for a1-A, 32.768-kHz sinusoidal input current.Although these results tend to supportthe measurement scheme, Williamscontends that it is worth noting thatTektronix measured the results. “Tektronixdoes not guarantee performancebelow the specified 3 dB, 25-kHz lowfrequencyroll-off. A3 and A4 contributea gain of 200, resulting in a total amplifiergain of 224,000. This figure resultsin a 1V/A scale factor at A4 referredto the CT-1’s output. A4’s LTC1563-232.7-kHz bandpass-filtered output feedsA5 through an LTC1968-based rms-to-dcconverter that provides the circuit’s outputs,”he says. The signal-processingpath, Williams explains, constitutes anextremely narrowband amplifier tuned tothe crystal’s frequency. Figure 3 depictstypical circuit waveforms. According toWilliams, the crystal drive at C1’s output(upper trace) causes a 530-nA rms crystalcurrent that the A4’s output (middletrace) and the rms-to-dc-converter input(lower trace) represent. “Peaking visiblein the middle trace’s unfiltered presentationderives from parasitic paths shuntingthe crystal,” he says.

Williams’ circuit provides several lessons. Measuring nanoamperes is difficult even when using integrating techniques. This problem was far more difficult, because he had to complete the measurement in real time. Further complicating matters was the fact that this ac measurement required a bandwidth of 32 kHz to capture the bulk of energy in the oscillator current waveform. Williams addressed these problems by using a sensor. The Tektronix CT-1 sensor (Reference 2) can cost as much as $500, but, without a good sensor, Williams would not have been able able to recover the signal from all the noise. In addition to good sensitivity, the CT-1 has a 50
output impedance that allows for lower noise-signalpaths than would a high-impedance output. Another important principlethat this example demonstrates is that itis essential to limit the bandwidth of thesignal path. By making a narrowbandamplifier chain, Williams discarded allthe noise contributions from frequenciesthat were not in his area of interest.Finally, Williams used good low-noisedesign principles in the circuit. Wiringcritical nodes in air minimizes leakagepaths, and the LT1028 is perhaps thelowest noise amplifier available fromany manufacturer when working from50Ω source impedance.

FEMTOAMPERE BIAS CURRENT
Paul Grohe, an application engineer at National Semiconductor, provides another remarkable example of measuring tiny currents. Years ago, National decided to sell the LMC6001, an amplifier that had a guaranteed bias current of 25 fA, implying that National needed to measure the bias current of each part to verify the specification. The test department could not accommodate test equipment in the setup; all the circuitry had to fit onto a standard probe card. Grohe and engineering colleague Bob Pease built a proof-of-concept fixture to demonstrate the feasibility of a small test circuit that could resolve to 1 fA (Figure 4). Many books and resources discuss using an integrating capacitor to measure small currents(Reference 3). The principle is that a small current can charge a small capacitorand that you can read that voltage toinfer the current. In some cases, the currentis an external current from a sensor.In this case, the current is leaving the amplifier-input pin. Figure 5 shows a simpletheoretical circuit in which the amplifieris measuring its own bias current.

The reality of measuring small currents is far more involved than the figure would suggest. First, Grohe could not use the part itself to measure its own bias current. If he had tried to use the part itself as the integrator, there would have been no way to calibrate the effects of a socket and other leakages associated with the test fixture. Doing so required a separate low-bias-current part as the integrator (Figure 6). Using a CMOS LMC660 amplifier ensured that the bias- current contribution would be less than 2 fA. By employing this technique, Grohe could simply remove any DUT (device under test), and the integrator would then have measured its own bias current as well as all the leakages from the test socket and the PCB on whichthe integrator was mounted.

Figure 7 shows that Grohe did not insert the DUT into a socket and that none of the pins are in contact with a PCB. To minimize leakage, Grohe brought up just two power pins as long, separate individual sockets that he did not mount to a PCB. Likewise, he hooked the pin to be tested to a socket and a 2-in. flying lead and connected that pin-and-socket combination to the integrating- amplifier input. To keep the DUT from running as an open loop, Grohe soldered together two sockets to bridge the output pins, which are suspended in air. Air currents can carry charged ions that can give false readings, so Grohe enclosed the entire DUT in ashielded copper-clad box.

The next issue was selecting an integrating capacitor. Initially, Grohe felt that the best capacitor would be an airdielectric capacitor, so he fashioned two large plates, measuring about 45 in., for the integrator capacitor. The size of this capacitor accounts for the size of the second copper-clad box on which the DUT box is mounted. Using a large capacitor proved to be a bad idea. The large area provides an ample target for cosmic radiation, creating ionized charges that interfere with the measurement (Figure 8). Grohe then minimizied the capacitor’s size while still using a good dielectric. It occurred to him that RG188 coax cable uses Teflon insulation. A 2-in. section of this cable provided the 10 pF for the integration capacitor (Figure 9). As a further benefit, the outside braiding would serve as shield. Grohe therefore hooked it to the low-impedance-output side of the amplifier. With the switch to this capacitor, the cosmic rays struck only once every 30 seconds or so. Grohe took the integrated measurement for 15 seconds and, by taking five measurements, negated their effect. He then discarded any single outlying measurement. Any ionizing radiation sources, even an old watch with a radium dial, can cause cosmicray problems. Note that Grohe pried up the input pin of the amplifier to preventleakage from the PCB.

Before taking a measurement, you need to reset the integrating capacitor to zero. Using a semiconductor switch is impractical, because of leakage currents and the 5- to 20-pF capacitance that most analog switches offer. That capacitance exhibits the varactor effect, as well; it changes with applied voltage, further complicating measurement. To minimize these problems, Grohe used a Coto-reed relay. Knowing that the coil might couple to the internal reed when the relay was open, he specified a relay with an electrostatic shield. Much to his dismay, there still was a large jump in the measurement when the relay opened due to charge injection. It turns out that you can also look at a reed relay as a transformer, with the reed assembly representing a single turn. This phenomenon explains the failure of the electrostatic shield to prevent the interference. Magnetic fields inducing voltages in the high-impedance side of thecircuit caused the charge injection. The relay does not open instantaneously,and the pulse needed to energize the coilmakes a significant current injection justbefore the relay opens. Grohe minimizedthis problem by characterizing the absoluteminimum voltage swing neededto operate the relay he had installed. Itturned out that the relay would pull inwith 3.2V and drop out with 2.7V. Heused a set of resistor taps on an LM317adjustable regulator to control the outputbetween these two values. By choosingnot to energize the relay with a full5V, he minimized the jump in the integratoroutput and made it repeatable. Hethen nulled out the jump by injecting asmall current into the second gain-stageamplifier.

The gain stages are two low-noise amplifiers—the LMV751 or perhaps a chopper amplifier, such as the LM2011, would be suitable. Grohe sent this gained-up signal to a digital scope, which could record data and subtract the slope of the calibration run from the test runs to give a valid measurement. Grohe used two LS123-style one-shot circuits—one to trigger the relay and another to provide a suitable and repeatable time delaythat triggered the digital scope.

Grohe also understood that good lownoise- design principles also include the power rails to the parts, so he chose not to power the relay or digital circuits from the same power he used for the integrator and DUT. He employed a handful of fixed and variable regulators to provide 5V for the DUT and integrator, 8V for the relay-drive circuit, and a separate5V for the digital circuits.

Using this circuit, Grohe was easily able to resolve 1 fA of current and found that most of the LMC6001 parts he tested had less than 5 fA of bias current, far exceeding the spec. He used this breadboard as the basis for a production-test circuit mounted on a standard probe card. (See references 4, 5, and 6 for more details about his design, includinga video of the system.)

Grohe would not use this circuit to measure femtoampere currents in his lab. “I would wheel out the Keithley 2400 electrometer,” he says. “We would have used that instrument to test the LMC6001 in manufacturing had the fab allowed us to use external test equipment.”His faith in Keithley is well-placed. On its Web site, the company offers—for free—an excellent article on measuringattoamperes (Reference 7), aswell as a book on delicate measurements(Reference 8).

Grohe and Pease’s integration approach is not limited to laboratory setups. Texas Instruments has created a line of parts that can measure in the femtoampere range and provide a digital output to boot. The line includes a single-channel DDC101 as well as the improved-sensitivity, dual-channel DDC112, which provides for external integrating capacitors. The four- and eight-channel DDC114 and DDC118 have a charge sensitivity of 12 pC (Reference 9). The sample rate for these 20-bit parts reaches 3 kHz. You must be knowledgeable aboutphysics to attempt these measurements.

If the DDC112 can measure 12 pC of charge and you want to measure 12 pA of current, you need to set the integrating time to 1 second, the maximum the DDC114 allows. It is impossible to obtain a 3-kHz update rate if the part’s integration interval is a full second. However, using the part configured in this fashion yields a 20-bit value at the end of the conversion. In other words, the DDC (direct digital converter) can resolve femtoampere currents, although at reduced accuracy. The input bias of the part is 20 fA, but your system’s software can calibrate out this value, so the part should still be able to resolve to very low levels. Bear in mind that this type of sensitivity makes it difficult to calibrate the system just once in the factory and then have it work for all time. As temperature increases, the bias current increases, doubling every 10C, and leakages as well as sensor drift can develop on your board. Providing the means for field calibration at power-up or more frequently is always a good idea when measuring currents in the femtoampere range. Texas Instruments offers evaluation boards for these parts that you can get up and running in hours, measuring currents too small for even a good handheld digital voltmeter(Figure 10).

According to Jim Todsen, product-linemanager for oversampling converters at TI and patent holder on the technologythat the part uses, the DDC line’s developmentstarted with the Burr-BrownACF2101—a dual switched integratorfront end that provides a single-chip optionfor the current-to-voltage function.The benefit of a dual integrator, Todsenexplains, is that it is always collectinginput current. While one integrator issampling the input, the other side is presentingits integrated value to the ADC,and this process continues for as longas you need measurements. “After theACF2101 converts the input currentto a voltage,” he says, “a discrete high resolution ADC digitizes it. The DDC112brought together both the current-tovoltagefunction of the ACF2101 andthe digitization of the high-resolutionADC in one chip.” He attributes thisachievement to advances in wafer processingthat allow high levels of mixedsignalintegration as well as TI’s developmentof a high-speed delta-sigma corethat can provide the required speed andresolution to measure the front-end signals.“In addition,” he notes, “we tookadvantage of having all the circuitelements under our control to optimizefor very-low-leakage inputs and very stableperformance over long integrationperiods.”

MORE AT EDN.COM

Go to www.edn.com/070426cs for this article’s associated vendor box or to post a comment on this article.

These applications should convince you of the difficulty of measuring small currents. They should also convince you of the value of using proven parts and equipment—whether Analog Devices’ AD549, National Semiconductor’s LMC660, TI’s DDC114 integrated circuits, Keithley’s 2400 parametermeasurement unit, or Agilent’s 4156parameter-measurement unit—in this d e m a n d i n gapplication.Remember,though, thatthese remarkableparts andinstrumentsare not magicboxes. Youcan take advantageof them only by removing noisesources and leakage paths from yourboard or test setup. Understanding opampspecifications for voltage and currentnoise will help you select the rightpart (Reference 10). In the meantime,if your boss wants to know why you need$5 or $10 for a chip or thousands of dollarsfor an electrometer, you can nowexplain that, with the challenges entailedin measuring small currents, thisequipment is a bargain.

REFERENCES

Long, James, “Sidebands be gone, or let there be (no) light,” EDN, Oct 12, 2006, pg 40, www.edn.com/article/ CA6378105. “AC current probes,” www.tek.com/ site/ps/0,,60-12572-INTRO_EN,00. html. Mancini, Ron, “The nuances of op-amp integrators,” EDN, March 18, 2004, pg 28, www.edn.com/article/ CA402150. Pease, Bob, “What’s All This Teflon Stuff, Anyhow?” Feb 14, 1991, www.national. com/rap/Story/0,1562,4,00.html. Pease, Bob, “What’s All This Femtoampere Stuff, Anyhow?” Sept 2, 1993, www.national.com/rap/Story/ 0,1562,5,00.html. www.national.com/nationaltv. Daire, Adam, “Counting Electrons: How to measure currents in the attoampere range,” Keithley Instruments Inc, September 2005, www.keithley.com/ data?asset=50390. www.keithley.com/wb/141. “Quad Current Input 20-Bit Analogto- Digital Converter,” Texas Instruments Inc, June 2005, http://focus.ti.com/lit/ ds/symlink/ddc114.pdf. Brisebois, Glen, “Op Amp Selection Guide for Optimum Noise Performance,” Linear Technology Design Note 355, January 2005, www.linear.com/pc/ downloadDocument.do?navId=H0,C1, C1154,C1009,C1021,P2440,D6539.